System and method for isolating the undriven voltage of a permanent magnet brushless motor for detection of rotor position

ABSTRACT

The system and method disclose for the controlling of sequential phase switching in driving a set of stator windings of a multi-phase sensorless brushless permanent magnet DC motor. A motor controller controls a power stage that drives two windings of a set of three windings in the motor with pulse width modulated signal. A plurality of voltage values on an undriven winding of the set of three windings are sampled within a window of time, wherein a period beginning when the driven windings are energized and ending when the driven windings are de-energized encompasses the window of time. The sampled voltage values are processed. When the processed voltage values exceed a threshold, the motor controller changes which two windings are driven.

CROSS-REFERENCE

This application is a Continuation in Part to U.S. application entitled“Circuit and Method for Sensorless Control of a Permanent MagnetBrushless Motor During Start-up,” having Ser. No. 13/800,327 filed Mar.13, 2013 and claims the benefit of U.S. Provisional Applicationentitled, “Circuit and Method for Sensorless Control of a BrushlessMotor During Start-up,” having Ser. No. 61/651,736, filed May 25, 2012,which is entirely incorporated herein by reference.

FIELD

The present disclosure is generally related to motor controllers, andmore particularly is related to a system and method for sensorlesscontrol of a permanent magnet brushless motor during start-up.

BACKGROUND

Sensored brushless motor technology is well-known and is useful forminimal flaw control at low speeds and reliable rotation. A sensoredsystem has one or more sensors that continuously communicate with amotor controller, indicated to it what position the rotor is in, howfast it is turning, and whether it is going forward or reverse. Sensorsin a sensored system increase cost and provide additional pieces thatcan break or wear down, adding durability and reliability issues.Sensorless systems can read pulses of current in the power connectionsto determine rotation and speed. Sensorless systems tend to be capableof controlling motors at higher speeds (e.g., revolutions per minute(“RPM”)), but may suffer jitters under a load at very low startingspeeds, resulting in a performance inferior to sensored brushlessmotors.

Jitter is a phenomenon that occurs with sensorless brushless motorsystems at initial starting speed and generally no longer exists afterthe motor has gained sufficient speed. Jitter comes about because at lowor zero speed, the sensorless algorithm does not have enough informationto decide which windings to energize and in what sequence. One commonsolution for starting a sensorless system is to energize one windingpair to lock the rotor at a known position. The motor windings are thencommutated at a pre-defined rate and PWM duty cycle until the rotorreaches a speed sufficiently high for the sensorless control to engage.However, even this solution will cause jitter during startup,particularly if there are time varying loads. Jitter can be decreased ormade imperceptible for loads with minimal initial torque or predictableinitial torque. However, some motor application/use situations (such asstarting an electric motor bike moving uphill) demand significant torquefor initiation, and the initial torque is highly unpredictable. Use ofsensorless brushless motor systems is sometimes discouraged forlow-speed high-torque maneuvers, like rock-crawling or intricate anddetailed track racing of an electric motor vehicle/bike, because in suchdifficult situations, significant jittering may occur and can lead topremature motor burnout.

FIG. 1 is a block diagram of a motor control system 10 in a three-phasepower stage, as is known in the prior art. Many three-phase motorcontrol systems 10 include a controller having a control signalgenerator 12, a gate driver 14, and a power stage 16. In case ofsensorless control, feedback circuits are also included, specifically adetection network 18 and a current sensing circuit 20, which utilizessense resistor R_(SENSE). In general, a goal of sensorless control is todetect a motor response to an applied pulse width modulated (PWM) sourcevoltage to identify rotor position and movement.

Similarly, a current sense circuit 20 may be used to detect themagnitude and direction of motor current across driven windings. Lowside shunt monitoring is used regularly. An often used configuration forlow side monitoring is shown in FIG. 1. One skilled in the art caneasily adopt alternative current sensing techniques such as monitoringphase current in each inverter branch including high-side monitoring andthis alternative technique is known to those having ordinary skill inthe art.

The control signal generator 12 is often powered from a low voltagesource. As a result, a function of the gate driver 14 includes shiftingthe low voltage control signals to levels that match input requirementsof the power stage 16. The power stage 16 includes semiconductorswitching devices. MOSFETs are shown in FIG. 1, but other devices suchas IGBTs may be used. The control signal can be made to generatetrapezoidal (a.k.a. block or 6 step commutation) or sinusoidal drivefrom the power source V_(pwr). Pulse width modulation is typically usedwith trapezoidal drive in brushless DC (BLDC) motor control. Systemsrequiring lower audible noise or lower torque ripple benefit fromsinusoidal drive.

Those skilled in the art with respect to PWM drive techniques understanda variety of modes to generate trapezoidal, sinusoidal, or othercontrol. The motor response to a PWM drive can be detected via voltageon the motor phases and/or phase current(s).

As shown in FIG. 1 for a brushless DC motor control, the power stage 16is driven such that current flows into a first motor phase (for example,phase U) and exits a second motor phase (for example, phase V). Therotor (not shown) position within the motor 30 dictates which phase pairto drive to attempt to achieve full torque and smooth (jitter-free)rotation of the rotor. The feedback controls are used to deduce rotorposition.

FIG. 2 is an illustration of a wye-connected motor 30, as is known inthe prior art. The wye-connected motor 30 in this illustration has asingle-pole pair permanent magnet rotor 32 positioned such that itssouth pole 34 is proximate to the winding of the U-phase 36. Under theseconditions, it is obvious to one skilled in the art that the W-phase 38and the V-phase 40 are the appropriate phase pair to drive in order toinitiate rotation of the rotor 32. The polarity of the permanent magnetrotor 32 determines the direction of current flow through the phase.Hence, the power stage 16 connects the W-phase 38 to V_(pwr) and theV-phase 40 to ground 24 resulting in current flow into the W-phase 38and exiting the V-phase 40, as represented with the current arrows. Anet effect of current flowing through coils W-phase 38 and V-phase 40 asshown in FIG. 2 is the formation of an electromagnet having a north poleat the W-phase 38 and a south pole at V-phase 40. This electromagnetproduces a repulsive force between permanent magnet N-pole 42 and theelectromagnet N-pole formed at the W-phase 38 and an attractive forcebetween permanent magnet N-pole 42 and the electromagnet S-pole formedat the V-phase 38.

As N- and S-poles are attracted to each other, if the electromagnetpersisted long enough in this current flow configuration, the resultingtorque will move the permanent magnet N-pole 42 to a position shortlyafter the V-phase 40 and the permanent magnet S-pole 34 to a positionshortly before the W-phase and rotation of the permanent magnet rotor 32would stop. To perpetuate rotation of the permanent magnet rotor 32, thepower stage 16 must commutate to a new phase pair. The optimumcommutation point is a function of the rotor position relative to thecoil of the undriven phase (the phase not driven by V_(pwr)). In FIG. 2,the U-phase 36 is the undriven phase. Ideally, the rotor angle wouldspan −30° to +30° with respect to alignment with the coil of theundriven phase. As this 60° span is one sixth of one electricalrevolution, it is commonly referred to as one sextant.

FIG. 3 is a 6-step commutation process further defined by Table 1, as isknown in the prior art. Given the conditions illustrated in FIG. 2, ahigh level description of the sequence of steps commonly referred to as6-step commutation process is outlined in Table 1 and furtherillustrated in FIG. 3.

TABLE 1 Six-step commutation sequence for a wye-connected motor shown inFIG. 2 Se- Driven quence Phase N-pole position S-pole position RotorStep Pair relative to phases relative to phases Angle 0 WV W + 30° to V− 30° U − 30° to U + 30° 1.25-1.75 1 WU V − 30° to V + 30° U + 30° to W− 30° 1.75-2.25 2 VU V + 30° to U − 30° W − 30° to W + 30° 2.25-2.75 3VW U − 30° to U + 30° W + 30° to V − 30° 2.75-0.25 4 UW U + 30° to W −30° V − 30° to V + 30° 0.25-0.75 5 UV W − 30° to W + 30° V + 30° to U −30° 0.75-1.25

The 6-step commutation sequence results in one electrical revolution.Given this simplified example, it is understood that a properly drivenpermanent magnet rotor will be driven one mechanical revolution whenthis six-step process is complete. An increase in number of pole pairresults in an equivalent increase in the number of electricalrevolutions per mechanical revolution. Comparing Table 1 and FIG. 2, itis understood that FIG. 2 illustrates Sequence Step 0 with the permanentmagnet N-pole 42 pushed from the W-phase 38 and pulled by the attractionto the V-phase 40. When the permanent magnet S-pole 34 reaches the U+30°position, the power stage 16 commutates to Sequence Step 1 drivingcurrent from the W-phase 38 to the U-phase 36 causing the U-phase tobecome the electromagnetic S-pole. Thus, the U-phase 36 repels or pushesthe permanent magnet S-pole 34 and the W-phase 38 attracts the S-pole,continuing the clockwise motion of the permanent magnet rotor 32.

Most current solutions to sensorless control of a brushless permanentmagnet motor utilize a symmetric pulse width modulation signal. FIG. 4Ais an illustration of one example of one symmetric pulse widthmodulation signal, as is known in the prior art. One cycle of asymmetric pulse width modulation signal may include a positive voltageV+ for a span of time T_(A) and then a negative voltage V− for the spanof time T_(B), where the absolute values of V+ and V− are equivalent anda full PWM period is T_(A)+T_(B). The span of time spent at V+ isreferenced as the energizing portion of the signal and the span of timespent at V− is referenced as the de-energized portion of the signal.FIG. 4B is an illustration of one example of one asymmetric pulse widthmodulation signal, as is known in the prior art. This signal includes aspan of time T₁ at V+ and a second span of time T₂ at approximately 0V.The sum of T₁ and T₂ represents the PWM period.

Others have developed solutions to sensorless control of a brushlessmotor during start-up as they relate to spindle motors for disk drives,compact disk (“CD”) drives, and digital video disk/digital versatiledisk (“DVD”) drives. However, CDs and DVDs do not offer significant orvarying resistance to rotor rotation during start-up. Because theinitial torque is predictable, performance parameters can bepre-programmed significantly. Other motors, such as those for electricscooters, can have varying resistance applied to the motor. Forinstance, the torque placed on a motor for an electric scooter may bedependent on the amount of weight on the electric scooter and whetherthe scooter is facing an incline, a decline, or on a level surface. Themotor characteristics are influenced by the magnitude of start-upcurrent needed to overcome the external torque on the motor. Theundriven phase is known to carry a voltage influenced by the current onthe driven windings and magnetic field from the rotor, among otherfactors, but has been regarded as too noisy and influenced by too manyfactors to provide useful information. The undriven phase could be asource of useful data if the noise on the winding could be filtered out.

Thus, a heretofore unaddressed need exists in the industry to addressthe aforementioned deficiencies and inadequacies.

SUMMARY

Embodiments of the present disclosure provide a system and method forsynchronizing sequential phase switching in driving a set of statorwindings of a multi-phase sensorless brushless motor. Briefly described,in architecture, one embodiment of the system, among others, can beimplemented as follows. The system discloses structure for controllingsequential phase switching in driving a set of stator windings of amulti-phase sensorless brushless permanent magnet DC motor. A motorcontroller controls a power stage that drives two windings of a set ofthree windings in the motor with pulse width modulated signal. Aplurality of voltage values on an undriven winding of the set of threewindings are sampled within a window of time, wherein a period beginningwhen the driven windings are energized and ending when the drivenwindings are de-energized encompasses the window of time. The sampledvoltage values are processed. When the processed voltage values exceed athreshold, the motor controller changes which two windings are driven.

The present disclosure can also be viewed as providing a method ofcontrolling motor switching. The method includes the steps of: driving apulse width modulated signal on two windings of a set of three windings;sampling a plurality of voltage values on an undriven winding of the setof three windings within a window of time, wherein a period beginningwhen the driven windings are energized and ending when the drivenwindings are de-energized encompasses the window of time; signalprocessing the sampled voltage values; and changing which two windingsare driven when the processed voltage values exceed a threshold.

Other systems, methods, features, and advantages of the presentdisclosure will be or become apparent to one with skill in the art uponexamination of the following drawings and detailed description. It isintended that all such additional systems, methods, features, andadvantages be included within this description, be within the scope ofthe present disclosure, and be protected by the accompanying claims.

BRIEF DESCRIPTION OF THE DRAWINGS

Many aspects of the disclosure can be better understood with referenceto the following drawings. The components in the drawings are notnecessarily to scale. Instead emphasis is being placed upon illustratingclearly the principles of the present disclosure. Moreover, in thedrawings, like reference numerals designate corresponding partsthroughout the several views.

FIG. 1 is a block diagram of a motor control system in a three-phasepower stage, as is known in the prior art.

FIG. 2 is an illustration of a wye-connected motor, as is known in theprior art.

FIG. 3 is a 6-step commutation process further defined by Table 1, as isknown in the prior art.

FIG. 4A is an illustration of one example of one symmetric pulse widthmodulation signal, as is known in the prior art.

FIG. 4B is an illustration of one example of one asymmetric pulse widthmodulation signal, as is known in the prior art.

FIG. 5 is a block diagram of a motor control system in a three-phasepower stage, in accordance with a first exemplary embodiment of thepresent disclosure.

FIG. 6 is an illustration of demodulated signals representing motorphases illustrated in FIG. 2.

FIG. 7 is an illustration of demodulated signals representing motorphases illustrated in FIG. 2 under the influence of high torque andcurrent.

FIG. 8 is an illustration of an exemplary voltage sense circuit that maybe used in conjunction with the motor control system in FIG. 5, inaccordance with the first exemplary embodiment of the presentdisclosure.

FIG. 9 is an illustration of an exemplary current sense circuit that maybe used in conjunction with the motor control system in FIG. 5, inaccordance with the first exemplary embodiment of the presentdisclosure.

FIG. 10A is an illustration of the demodulated undriven phase signalassociated with the motor control system of FIG. 5 driven with asymmetrical pulse width modulation signal, in accordance with the firstexemplary embodiment of the present disclosure.

FIG. 10B is an illustration of less than two periods of the undrivenphase signal associated with the motor control system of FIG. 5 drivenwith a symmetrical pulse width modulation signal, in accordance with theexemplary embodiment of the present disclosure.

FIG. 11A is an illustration of the demodulated undriven phase signalassociated with the motor control system when driven of FIG. 5 with anasymmetric PWM signal, in accordance with the first exemplary embodimentof the present disclosure.

FIG. 11B is an illustration of less than two periods of the demodulatedundriven phase signal associated with the motor control system driven ofFIG. 5 with an asymmetrical pulse width modulation signal, in accordancewith the exemplary embodiment of the present disclosure.

FIG. 12 is an illustration of a flowchart illustrating a method of usingthe motor control system 110 of FIG. 5, in accordance with the firstexemplary embodiment of the present disclosure.

DETAILED DESCRIPTION

FIG. 5 is a block diagram of a motor control system 110 for athree-phase power stage 116 for a sensorless, brushless permanent magnetDC motor 30, in accordance with a first exemplary embodiment of thepresent disclosure. The motor control system 110 includes a controllerunit 160 having a control signal generator 112, a memory device 162, aprocessing unit 164, a signal acquisition device 166, and ananalog-to-digital converter 170. The control signal generator 112 feedssix inputs into a gate driver 114. The gate driver 114, which may bepowered by an independent power source (not illustrated), controls sixMOSFET switches 168 in the power stage 116. Manipulation of the switchesdetermines current flow from the power source V_(pwr) through the statorwindings 36, 38, 40 in the motor 30.

The voltage sense circuit 118 and current sense circuit 120 are used forclosed loop control of the motor. The power stage 116 has 6 switchesgrouped in pairs. Each switch pair is configured as a half bridge. Eachswitch has a control input. The outputs of power stage 116 are fed intothe 3-phase BLDC motor windings U 36, V 40, W 38. The power stage 116 issupplied by a voltage source V_(pwr) having a DC voltage, which thepower stage uses to supply a pulse width modulation signal to thewindings U 36, V 40, W 38. The current return path for the voltagesource V_(pwr) is through ground via current sense resistor R_(SENSE).The power stage 116 for a trapezoidally controlled pulse width modulatedbrushless DC motor 30 typically energizes two motor windings of the setof three windings 36, 38, 40 at a time.

A voltage signal is available at the undriven phase. This voltage signalcan be used to generate a commutation signal by demodulating theundriven phase voltage synchronously with the PWM switching rate. Thecommutation signal, when a near-zero drive current is present, has aperiodicity of ½ electrical revolutions. The shape of this commutationsignal is related to the action of the permanent magnet rotor 32 on thestator windings 36, 38, 40. Demodulation can be performed by simplytaking the difference in voltage between the undriven phase and theswitching in two different driven states of the PWM when the PWM signalis symmetric. Demodulation can be performed by simply taking thedifference in voltage between the undriven phase and a reference voltageduring the energizing portion of the PWM when the PWM signal isasymmetric. When a materially-greater-than-zero current is driven intothe active pair of terminals, the signal has an added component with aperiodicity of a full electrical cycle.

FIG. 6 is an illustration of demodulated undriven phase signalsrepresenting motor phases illustrated in FIG. 2 and FIG. 3. Thesubscript D indicates the signal is from a demodulated undriven winding.Here, the undriven phase signals are illustrated for a ½ electricalcycle superimposed upon each other relative to rotor angle. The propercommutation time can be determined by monitoring the undriven phasesignal, derived from the demodulated undriven phase signal, andcommutating at a value that is a function of the motor current. As thecurrent increases, a comparison value will change, but FIG. 6 isrepresentative of a near-zero current through the driven windings.

As illustrated in FIG. 6, the dotted line U_(D) represents thedemodulated signal produced when the U-phase 36 is disconnected duringcommutation sequence step 0, and the WV phases 38, 40 are driven with aPWM wave. This drive combination is the connection that generates themost torque from the rotation position 1.25 to the 1.75 point on thex-axis, which is the sextant position

If the motor is being driven with torque pushing to the right, when 1.75point is reached, the motor is rotating in the proper direction, andcommutation from WV phases to WU phases should occur at the 1.75 point.Likewise, if the rotor is rotating counterclockwise while beingelectrically driven clockwise, such as starting an electric scooter on ahill, U_(D) has negative slope between 1.25 and 1.75. If the 1.25 pointis reached, the prior commutation phase UV or commutation sequence step5 should be switched in. These points are associated with thedemodulated signal U_(D), reaching approximately 1.5 or −1.5 volts forforward or reverse commutation respectively, illustrated as THRESHOLD inFIG. 6. When 1.75 on the x-axis is reached, upon commutating to WUphases, the demodulated signal associated with the V-phase 40, i.e.V_(D), will then be generated. If 1.25 is reached (forced in reverse),the demodulated signal associated with the W phase, i.e. W_(D), willthen be generated.

If the commutation signal component from the permanent magnets isdominant, determining the time for commutation is straightforward. Thecommutation signal from the undriven phase is derived, and whenpre-determined values are reached, the motor is advanced to the next orprior phase. The prior phase advance is important, as the load may berotating in the direction opposite to the desired rotation upon start.For maximum torque, it is important that the commutation levels berelatively accurate.

When the required starting torque is high, amaterially-greater-than-zero current is needed through the drivenwindings to generate the high torque. The commutation breakpoint isharder to determine from the undriven phase signal when the drivenwinding current is high. The commutation signal transforms substantivelywith respect to rotational position when the current has surpassed anear-zero level.

FIG. 7 is an illustration of demodulated signals representing motorphases illustrated in FIG. 2 under the influence of high torque andcurrent. The proper values for the demodulated undriven winding (U_(D))signal at the previously identified commutation breakpoints (when therotor angle is 1.25 and 1.75) are 0V and 3V. Thus, if the motorcontroller operated with a threshold of −1.5V and 1.5V for commutation,as was shown in FIG. 6, the motor would not be able to obtain themaximum available torque that is acquired with proper motor commutation.In the forward motion case, the commutation will be too early, causing atransfer to a commutation sequence step that will provide less torque.In the case the motor is rolling backward, the commutation may be toolate to achieve high torque from the previous commutation step. Further,at slightly higher currents, the result may be failure to commutatealtogether leading the controller to errantly attempt to drive the motorin the wrong direction. The effect of current on the demodulated signalmay be different for even and odd sextants (commutation sequence steps0, 2, and 4 as opposed to 1, 3, and 5). Motor characteristics indicatethe portion of the demodulated signal associated with even sextantsvaries proportionally with current and the portion of demodulated signalassociated with odd sextants varies inversely with current.

FIG. 8 is an exemplary voltage sense circuit 118 that may be used inconjunction with the motor control system 110 in FIG. 5, in accordancewith the exemplary embodiment of the present disclosure. The voltagesense circuit 118 is placed in the feedback path of a first controlloop, between the power stage outputs 116 and the controller unit signalacquisition device 166. The voltage sense circuit 118 includes aresistor network comprising resistors R1, R2, R3, R4, and R5 coupledtogether as shown in FIG. 8. Voltage sense circuit 118 has three inputsconnected to three motor terminals, U 36, V 40, W 38. The voltage sensecircuit 118 superposes motor voltage response from each phase 36, 38, 40and divides the result to level in accordance with input requirementsfrom signal acquisition 166. The result includes the voltage on theundriven phase. While similar motor control configurations includevoltage sense circuits 118, these circuits are directed to retrieving aback EMF signal and regularly filtering out the undriven phase voltageto get a cleaner back EMF signal.

FIG. 9 is an exemplary current sense circuit 120 that may be used inconjunction with the motor control system 110 in FIG. 5, in accordancewith the exemplary embodiment of the present disclosure. The currentsense circuit 120 is placed in the feedback path of a second controlloop, between a current sense resistor R_(SENSE) and the controller unitsignal acquisition device 166. The power supply voltage levels ofcurrent sense circuit 120 and controller unit 160 are approximately thesame. Current sense circuit 120 includes an amplifier 174 configured fordifferential measurement of voltage across R_(SENSE), as shown in FIG.5. The amplifier 174 input common-mode voltage and gain are set suchthat amplifier output is at approximately mid-supply to facilitatemonitoring of R_(SENSE) current flowing in positive and negativedirection.

The motor control system 110 may be used to control a motor 30, such asthe motor 30 illustrated in FIG. 2. FIG. 10A is an illustration of thedemodulated undriven phase signal associated with the motor controlsystem 110 driven with a symmetrical pulse width modulation signal, inaccordance with the exemplary embodiment of the present disclosure.Signals V and W are driven signals on two terminals of the motor 30.FIG. 10A illustrates a 50% duty cycle PWM with complementary drive. Thedrive phase voltage will normally be a value between ground and thepower supply voltage V_(pwr). Typical switching frequencies are in therange of 1 kHz to 25 Khz, depending on motor size and construction aswell as other factors. The signal at the undriven phase is shown in FIG.10 as signal U. Signal U changes as a function of rotor position whichvaries the magnetic fields in the stator. The demodulated undriven phasesignal U_(D), which is used for position sensing, is derived bymeasuring the voltage difference on signal U between the high b_(n) andlow a_(n) level. This voltage difference can be viewed as demodulationof the position signal from the PWM signal. The demodulated signal iscompared with an established threshold, such as the threshold shown inFIG. 5, and used to determine the commutation breakpoint where the powerstage output will switch to a next winding pair to drive. Theillustration of U_(D) in FIG. 10 is analogous to the 1.25-1.75 rotorangle portion of the U_(D) curve in FIG. 5 operating with steady rotormovement.

FIG. 10B is an illustration of less than two periods of the demodulatedundriven phase signal associated with the motor control system 110driven with a symmetrical pulse width modulation signal, in accordancewith the exemplary embodiment of the present disclosure. The U and U_(D)curves in FIG. 10B are comparable to those curves in FIG. 10A. As isevident in FIG. 10B, the U curve is more volatile in the moment themotor switches between an energized portion of the PWM signal and ade-energized portion of the PWM signal and then stabilizes. The voltageon the undriven phase U while the motor is changing between energizedand de-energized portions of the PWM period is not as useful fordefining the rotor position or commutation points as it is when the Usignal has stabilized. As is shown in FIG. 10B, a number of samplesa_(n) may be taken during the energized portion and a number of samplesb_(n) may be taken during the de-energized portion. These samples may beaveraged or otherwise filtered to arrive at an overall value for eachportion. The difference of the filtered a_(n) and filtered b_(n)provides a single demodulated voltage value for that PWM period as shownin FIG. 10B.

FIG. 11A is an illustration of the demodulated undriven phase signalassociated with the motor control system 110 when driven with anasymmetric PWM signal, in accordance with the first exemplary embodimentof the present disclosure. Signals V and W are driven signals on twoterminals of the motor 30. When the motor is operating in step 0 of thecommutation sequence as shown in FIG. 3, the W+ gate and the V− gatewill close when energizing while the other four gates are open. Whilede-energizing, the W− gate and the V− gate will close, disconnecting theset of windings from V_(pwr) and connecting the W and V phases to eachother and ground. The drive phase voltage will normally be a valuebetween ground and the power supply voltage. Typical switchingfrequencies are in the range of 1 KHz to 25 KHz, depending on motor sizeand construction as well as other factors. The signal at the undrivenphase is shown in FIG. 11A as signal U. Signal U changes as a functionof rotor position which varies the magnetic fields in the stator. Thedemodulated undriven phase signal, U_(D), which is used for positionsensing, is derived by measuring the voltage during the energizing phasewith respect to a reference voltage. This comparison of the measuredvoltage with a reference voltage is at least part of the demodulationstep. This demodulated signal is compared with an established threshold,such as the threshold shown in FIG. 6, and used to determine thecommutation breakpoint where the power stage output will switch to anext winding pair to drive. The illustration of U_(D) in FIG. 11 isanalogous to the 1.25-1.75 rotor angle portion of the U_(D) curve inFIG. 6 with a rotor rotating at a constant speed.

FIG. 11B is an illustration of less than two periods of the demodulatedundriven phase signal associated with the motor control system 110driven with an asymmetrical pulse width modulation signal, in accordancewith the exemplary embodiment of the present disclosure. The U and U_(D)curves in FIG. 11B are comparable to those curves in FIG. 11A. As isevident in FIG. 11B, the U curve is more volatile in the moment themotor switches between an energized portion of the PWM signal and ade-energized portion of the PWM signal and then stabilizes. The voltageon the undriven phase U while the motor is changing between energizedand de-energized portions of the PWM period is not as useful fordefining the rotor position or commutation points as it is when the Usignal has stabilized. As is shown in FIG. 10B, a number of samplesa_(n) may be taken during the energized portion. Samples may be takenduring the de-energized phase but are not required for practice of thedisclosed invention. These samples may be averaged or otherwise filteredto arrive at an overall value for each portion. The difference of thefiltered a, and a reference voltage provides a single demodulatedvoltage value for that PWM period as shown in FIG. 11B.

FIG. 12 is an illustration of a flowchart illustrating a method of usingthe motor control system 110 of FIG. 5, in accordance with the exemplaryembodiment of the present disclosure. It should be noted that anyprocess descriptions or blocks in flow charts should be understood asrepresenting modules, segments, portions of code, or steps that includeone or more instructions for implementing specific logical functions inthe process, and alternate implementations are included within the scopeof the present disclosure in which functions may be executed out oforder from that shown or discussed, including substantially concurrentlyor in reverse order, depending on the functionality involved, as wouldbe understood by those reasonably skilled in the art of the presentdisclosure.

As is shown by block 202, a pulse width modulated signal is driven ontwo windings of a set of three windings. A plurality of voltage valuesare sampled on an undriven winding of the set of three windings within awindow of time, wherein a period beginning when the driven windings areenergized and ending when the driven windings are de-energizedencompasses the window (block 204). The sampled voltage values areprocessed (block 206). Two different windings are driven when theprocessed voltage values exceed a threshold (block 208).

The step of changing which two windings are driven may involve changingwhich phases are driven after the demodulated measured voltage hasexceeded the threshold for a set period of time. The undriven voltagesignal may experience noise, and that noise may cause the threshold tobe surpassed prematurely and temporarily. Verifying that the demodulatedmeasured voltage continues to exceed the threshold for a period of timediminishes the possibility that the threshold is surpassed as a resultof noise instead of properly identified rotor position.

The threshold may be set as a function of the pulse width modulatedsignal. For instance, as an amplitude of the pulse width modulationsignal increases, the absolute value of the thresholds should increaseto properly compensate for the undriven winding voltage also increasingin value. The threshold may be predetermined and modified as a functionof a characteristic of the pulse width modulated signal. Similarly, thedemodulated measured voltage value may be modified within the motorcontroller as a function of the pulse width modulated signal to allowthe demodulated measured voltage value to intersect the threshold at theproper rotor rotation angles.

The demodulated measured voltage may be modified by scaling thedemodulated measured voltage.

The signal processing step may be performed by a signal processingcircuit containing at least one delta-sigma analog to digital converter170. The delta-sigma analog to digital converter may have a sample rateof at least sixteen (16) times the PWM frequency. The signal processingstep may further include at least two analog to digital converters. Ananalog summing network may obtain the processed voltage values furthercalculated by measuring a difference between the undriven winding and anaverage of the two driven windings for an analog to digital converterinput.

As is illustrated in FIG. 10B and FIG. 11B, some of the data in theundriven voltage signal is more desirable than other data. Thus, datacollection may be a sampling of undriven voltage values at discretetimes relative to the switching between energized and de-energizedstates of the motor. The time from which sampling begins until itconcludes during one energized or de-energized portion is termed a“window” herein. An exemplary window 190 is illustrated in FIGS. 10B and11B. The first sample taken and utilized in a calculation of ademodulated undriven voltage symbolizes the opening of the window andthe last sample within the window symbolizes the closing of the window.A set of switches in a power stage command the pulse width modulatedsignal on the driven windings. The window may open at the moment whentwo switches in the power stage close, and the window may close at themoment when at least five of the switches in the power stage are open.At those switching moments, it can be seen from the illustrations thatthe undriven voltage ripples. The system may be designed to manipulatethe window to collect undriven voltage data that is less impacted by theripples, thereby isolating a portion of the voltage on the undrivenwinding more directly attributable to rotor position.

In one variant embodiment, the window may open a delayed time frameafter two switches in the power stage close, and the window may close atthe moment when at least five of the switches in the power stage areopen. The delayed time frame may be until the processor determines anundriven voltage value exceeding 50% of the voltage supply V_(pwr). Fornegative voltages, only the magnitude of the voltage may be consideredto determine if 50% of the voltage supply V_(pwr) is exceeded.

More complex methods may be used to define the delayed time frame. Theprocessor may calculate a deviation of the plurality of voltage valuessampled, and the window spans a set of consecutive sampled voltagevalues that vary less than a threshold percentage from a mean of thevoltage values, represented by:|X _(i) −X _(ave) |/X _(ave) <K.where K is a threshold deviation constant, X_(ave) is mean of thedemodulated voltage values over the window and X_(i) is each demodulatedsample voltage values in the window series.

The processor may sample the undriven voltage across the duration of anenergized or de-energized portion and then define the window after themotor proceeds to the next portion. The sampled undriven voltage thatwas not within the window may be discarded by the processor whencalculating the demodulated voltage. The processor may determine thewindow is closed when the motor proceeds to the next portion and thendefine the opening of the window as a determined time frame before thewindow closed. For example, once the window may be termed open 0.5 msecbefore the window closed or it may be opened such that the window spans50% of the energized/de-energized portion. The determined time frame maybe a function of duty cycle of the pulse width modulation signal.

The values within the windows 190 can be processed in a number ofdifferent ways to achieve a value representative of the undriven voltageduring the relative energized/de-energized portion. As mentioned, thesampled voltage values within the window may be averaged. The sampledvoltage values may each be multiplied by a weighting function andsummed. The weighting function may be a time-varying periodic functionsynchronous with the PWM signal. Each of the sampled voltage values maybe multiplied by a weighting function, and a subset of the previousfilter outputs may be multiplied by a weighting function and theproducts summed together, a process associated with infinite impulseresponse filters to those having ordinary skill in the art. The step ofsignal processing may be initiated prior to the step of sampling.

When the pulse width modulated signal has a 50% duty cycle, theplurality of voltages may be multiplied by a filter with a sine wave anda cosine wave weighting function. The sine wave and cosine wave filteris multiplied by the plurality of sampled voltage values and summed, andthe window is defined by a period in which the sine wave filter sum isapproximately zero.

FIRST EXEMPLARY COMMUTATION BREAKPOINT CALCULATION

A pulse width modulation signal is provided to two windings at a levelthat provides a near zero average current (I_(min)) over the twowindings. A first set of voltage data representing the motor voltageresponse signal on the undriven phase 36, spanning at least an entiresextant, is obtained. A first set of current data representing thedriven phase current is collected corresponding to each data point inthe first set of undriven voltage data. The process is repeated with apulse width modulation signal that provides a mid-level drive phasecurrent (a.k.a. I_(mid)) and again with a pulse width modulation signalthat provides an approximately maximum drive phase current (a.k.a.I_(max)).

A first set of coefficients representing the influence of mid-levelvalues of current is calculated based on first and second current datasets.COeff_(midCurrent)=(V _(MTR)(I _(mid))−V _(MTR)(I _(min)))/(I _(mid) −I_(min))

Where V_(MTR) is the demodulated motor voltage response signal based onthe undriven phase 36.

A second set of coefficients representing the influence of max-levelvalues of current is calculated based on first and third current datasetsCoeff_(maxCurrent)=(V _(MTR)(I _(max))−V _(MTR)(I _(min)))/(I _(max) −I_(min))

The effect of current on the commutation signal is different in oddsextants compared to even sextants. Therefore, said first and secondsets of coefficients are created for both even and odd sextants.

Coeff_(midCurrent) (odd)

Coeff_(midCurrent) (even)

Coeff_(maxCurrent) (odd)

Coeff_(maxCurrent) (even)

The resultant coefficient values can be used as-is under specificconditions. For example, if an application runs at specific currentsbecause the motor drives known loads, then the coefficients can bestored in a lookup table. At each operating current level, thecoefficients can then be read from the table and used to compensate theundriven phase signal for that current.

Another method of modifying the threshold and/or demodulated voltageincludes transforming the resultant coefficient values into Slope andIntercept values for even and odd sextants, which can then be generallyapplied for a wide set of current values. The Slope and Intercept valuesare stored in memory.

The coefficient as a function of current is calculated as:Coefficient(I)=slope*I _(avg)+intercept

In this equation, I_(avg) is the average driven phase current, obtainedin this example via amplifier 174 in difference configuration monitoringlow side shunt resistor and generally described as current sense blockin FIG. 4 and FIG. 9. The amplifier output is sampled and digitized inboth the on and off portions of the PWM cycle. The values are digitallyprocessed to produce the average motor phase current in the PWM cycle.The Slope and Intercept values may be obtained from memory device 162.Sextant parity determines whether Slope and Intercept data for odd oreven sextants is used.

Slope is effectively calculated as ΔV/ΔI, hence, Coefficient(I) hasunits of resistance.

A correction factor as a function of current is then calculated as:V _(CF)(I)=I _(avg)*Coefficient(I)

Controller unit memory device 162 contains constant values representingmotor characteristics. Constant value(s) for commutation breakpoint isstored in memory device 162. Slope and intercept values are stored inmemory device 162.

Processing unit 164 performs arithmetic calculations based on stored andmeasured data. Specifically, the correction factor, V_(CF)(I), iscalculated and the motor voltage response on the undriven phase isdemodulated. The processing unit 164 inverts the polarity of thedemodulated signal in every other sextant such that the slope of thedemodulated signal with respect to the direction of the applied torqueis positively independent of the sextant. The processing unit 164modifies the demodulated signal with the correction factor in accordancewith the winding current. The processing unit 164 calculates directionof the demodulated signal based on its slope between commutationbreakpoints, thereby confirming direction of rotation. A differencebetween first and second demodulated signal data points taken betweenconsecutive commutation breakpoints is compared to a threshold value. Adifference value greater than the threshold value indicates positiveslope, while a difference value less than the threshold value indicatesnegative slope. The definition of slope by way of comparison to athreshold value is arbitrary. For example, a difference value less thana threshold value could just as well define a positive slope.

The processing unit 164 compares a modified/corrected demodulated signalto a stored forward commutation breakpoint. At least one occurrence ofthe combination of a modified demodulated signal having value greaterthan the forward commutation breakpoint value and confirmed forwarddirection of rotation results in processing unit 164 controlling thecontrol signal 112 to commutate the power stage 116 to a next phasepair. Requiring multiple occurrences of the satisfying condition priorto commutating may increase system robustness. The processing unit 164compares a modified/corrected demodulated signal to a stored reversecommutation breakpoint. At least one occurrence of the combination of amodified demodulated signal having value less than the reversecommutation breakpoint value and confirmed reverse direction of rotationresults in processing unit 164 controlling PWM 112 to commutate thepower stage 116 to a previous phase pair. Requiring multiple occurrencesof the satisfying condition prior to commutating may increase systemrobustness.

An average current across the driven windings can be acquired a numberof ways, including measurement and modeling, some of which are known tothose skilled in the art. One useful method for obtaining the currentacross the driven windings is averaging a current measured by an analogto digital convertor and a current sense mechanism. As is discussedabove, the average current is used to modify at least one of thethresholds and the demodulated measured voltage.

When the rotor rotates fast enough, relative to other motorcharacteristics and operating conditions, a reliable back EMF signalbecomes available. Use of a reliable back EMF signal to controlcommutation from driven pair to driven pair is well known in the art.Thus, the techniques disclosed herein are designed for controllingcommutation when the rotor is not moving or is rotating at speeds belowwhich a reliable back EMF signal is available. The motor controlswitches to the back EMF commutation technique when a rotational speedof the rotor surpasses a speed threshold such that the reliable back EMFsignal is available.

It should be emphasized that the above-described embodiments of thepresent disclosure, particularly, any “preferred” embodiments, aremerely possible examples of implementations, merely set forth for aclear understanding of the principles of the disclosed system andmethod. Many variations and modifications may be made to theabove-described embodiments of the disclosure without departingsubstantially from the spirit and principles of the disclosure. All suchmodifications and variations are intended to be included herein withinthe scope of this disclosure and protected by the following claims.

What is claimed is:
 1. Method of controlling motor switching, the method comprising: manipulating a gate controller with a control unit to drive a pulse width modulated signal on two windings of a set of three windings within a motor; sampling a plurality of voltage values on an undriven winding of the set of three windings within the motor within a window of time with a voltage sense circuit, wherein a period beginning when the driven windings are energized and ending when the driven windings are de-energized encompasses the window of time; signal processing the sampled voltage values from the voltage sense circuit, wherein the signal processing is completed within the controller unit; and changing which two windings are driven when the processed voltage values exceed a threshold with the controller unit manipulating the gate controller.
 2. The method of claim 1, wherein a set of switches in a power stage control the pulse width modulated signal on the driven windings, wherein the window of time begins when two switches in the power stage close and the window of time ends when at least five of the switches in the power stage are open.
 3. The method of claim 1, wherein a set of switches in a power stage control the pulse width modulated signal on the driven windings, wherein the window of time begins a delayed time frame after two switches in the power stage close and the window of time ends when at least five of the switches in the power stage are open.
 4. The method of claim 3, further comprising identifying a first voltage value exceeding 50% of the pulse width modulated supply and the window of time begins when the first voltage value is sampled.
 5. The method of claim 3, further comprising determining a deviation of the plurality of voltage values and having the window of time span the consecutive sampled voltage values that vary less than a threshold percentage from a mean of the voltage values, thereby isolating a portion of the voltage on the undriven winding more directly attributable to rotor position.
 6. The method of claim 1, wherein a set of switches in a power stage control the pulse width modulated signal on the driven windings, wherein the window of time ends when at least five of the switches in the power stage are open and the window of time is then defined as beginning a determined time frame before the window of time ends.
 7. The method of claim 6, wherein the time frame is determined as a function of the duty cycle.
 8. A method of controlling motor switching, the method comprising: manipulating a gate controller with a control unit to drive a pulse width modulated signal on two windings of a set of three windings within a motor; sampling a first plurality of voltage values on an undriven winding of the set of three windings within the motor within a first window of time with a voltage sense circuit, wherein a period beginning when the driven windings are energized and ending when the driven windings are de-energized encompasses the first window of time; sampling a second plurality of voltage values on the undriven winding of the set of three windings within the motor within a second window of time with the voltage sense circuit, wherein a period beginning when the driven windings are de-energized and ending when the driven windings are energized encompasses the second window of time; signal processing the first sampled voltage values and the second voltage values from the voltage sense circuit, wherein the signal processing is completed within the controller unit; and changing which two windings are driven each time the processed voltage values surpass a threshold with the controller unit manipulating the gate controller.
 9. The method of claim 8, wherein the first plurality of voltage values are processed into a first filtered voltage value and the second plurality of voltage values are processed into a second filtered voltage value and the processed voltage value is a difference of the first filtered voltage value and the second filtered voltage value.
 10. The method of claim 8, wherein the signal processing the sampled voltages further comprises multiplying each of the sampled voltages by a weighting function and summing the products.
 11. The method of claim 10, wherein the weighting function is a time-varying periodic function synchronous with the PWM signal.
 12. The method of claim 8, wherein the signal processing the sampled voltages further comprises multiplying each of the sampled voltages by a weighting function and multiplying a subset of the previous signal processing output values by a function and summing the products.
 13. The method of claim 8, wherein the signal processing is initiated prior to the step of sampling.
 14. The method of claim 8, wherein the pulse width modulated signal has a 50% duty cycle and the plurality of voltages are filtered with a sine wave and a cosine wave, wherein the sine wave filter is multiplied by the plurality of voltages and summed, and wherein the window defines a period in which the sine wave filter sum is zero.
 15. The method of claim 8, wherein the weighting function is a time-varying periodic function synchronous with the PWM signal.
 16. The method of claim 8, wherein the signal processing step is performed by a signal processing circuit containing at least one delta-sigma analog to digital converter.
 17. The method of claim 16, wherein the at least one delta-sigma analog to digital converter has a sample rate of at least sixteen times the PWM frequency.
 18. The method of claim 8, wherein the signal processing further comprises utilizing at least two analog to digital converters.
 19. The method of claim 8, wherein an analog summing network obtains the processed voltage values further calculated by measuring a difference between the undriven winding and an average of the two driven windings for an analog to digital converter input.
 20. The method of claim 8, wherein the motor is a three-phase permanent magnet motor.
 21. A system for controlling motor switching in a sensorless BLDC motor having a set of three stator windings, the system comprising: a controller unit comprising a control signal generator, a memory device, a processing unit, and a signal acquisition device; a power stage having a plurality of switches in communication with the control signal generator, wherein the power stage receives a control signal from the control signal generator and a power signal from a power source, wherein the power stage drives two windings of the set of three stator windings with a pulse width modulation signal and leaves one stator of the three stator windings undriven; wherein the signal acquisition device in communication with a voltage sense circuit receives a plurality of samples of a voltage on the undriven winding during an energized state of each pulse width modulation cycle, whereby a time interval for acquiring the plurality of samples is a window of time; wherein the processing unit in communication with the signal acquisition device demodulates the voltage on the undriven winding; and wherein the processing unit in communication with the power stage through the control signal generator and in communication with the memory device communicates with the power stage to change which two windings of the three stator windings are driven when the demodulated measured voltage surpasses a threshold stored on the memory device.
 22. The system of claim 21, wherein the window of time spans a portion of the undriven voltage during an approximate second half of the energized state of the pulse width modulation signal, thereby isolating a portion of the voltage on the undriven winding more directly attributable to rotor position.
 23. The system of claim 21, wherein the processing unit determines a deviation of the plurality of samples and defines the window of time as spanning a set of consecutive sampled voltages of the samples that vary less than a threshold percentage from a mean of the samples.
 24. A system for controlling motor switching in a sensorless BLDC motor having a set of three stator windings, the system comprising: a controller unit comprising a control signal generator, a memory device, a processing unit, a signal acquisition device, and an analog-to-digital converter; a power stage having a plurality of switches and in communication with the control signal generator and a power source, wherein the power stage receives a control signal from the control signal generator and the power signal from the power source, wherein the power stage drives two windings of the set of three stator windings within the motor with a pulse width modulation signal and leaves one stator of the three stator windings within the motor undriven; wherein the signal acquisition device in communication with the analog-to-digital convertor and a voltage sense circuit receives a first plurality of samples of a voltage on the undriven winding during an energized state of each pulse width modulation cycle, whereby a time interval for acquiring the plurality of samples is a first window of time; wherein the signal acquisition device receives a second plurality of samples of a voltage on the undriven winding during a de-energized state of each pulse width modulation cycle, whereby a time interval for acquiring the plurality of samples is a second window of time; wherein the processing unit in communication with the signal acquisition device demodulates the voltage on the undriven winding at least by calculating a difference between the first plurality of samples and the second plurality of samples; and wherein the processing unit in communication with the power stage through the control signal generator and in communication with the memory device communicates with the power stage to change which two windings of the three stator windings within the motor are driven when the demodulated measured voltage surpasses a threshold stored in the memory device.
 25. The system of claim 24, wherein the controller unit filters the first plurality of samples and the second plurality of samples with at least one delta-sigma analog to digital converter.
 26. The system of claim 24, wherein the at least one delta-sigma analog to digital converter has a sample rate of at least sixteen times the PWM frequency. 